Inductive power transfer pick up circuits

ABSTRACT

An inductive power transfer (IPT) pick-up circuit for receiving power from a primary conductor has a pick-up coil (L 2 ) and a compensation capacitor (C 2 ) so that the pick-up coil (L 2 ) may be resonant at the system operating frequency a switch (S 1 , S 2 ), and a plurality of reactive elements (L 3 , C 3 ) whereby when the switch is in one of an on state or an off state the additional reactive elements (L 3 , C 3 ) are resonant at the operating frequency to reduce power being supplied to an output of the pick-up circuit.

FIELD OF THE INVENTION

This invention relates to inductive power transfer (IPT) systems.

BACKGROUND

IPT pick-up circuits may process alternating current (AC) to providepower to a load. As the power and voltage output requirements of ACprocessing circuits suitable for IPT applications such as lightingrises, so does the strain on the AC switch used to regulate the power.In the standard AC processing pickup described in patent publication WO2010/030195A1, the VA product which the switch must be rated for risesapproximately in proportion to the square of the load dependent qualityfactor (i.e. Q₂ ²). The power capacity of the pickup, however, onlyrises in proportion to Q₂. Thus to achieve large power outputs andrelatively high circuit Q, a modified controller is required.

The switch and current ratings in the parallel tuned AC controllercircuits described in WO 2010/030195 have ratings that depend on thetopology and the amount of tuning required to drive the load. In veryhigh power applications the switch rating can be prohibitive andexpensive. For example, the parallel AC processing pickup circuit ofFIG. 1 must be rated to handle the peak resonant inductor current alongwith the peak load voltage, {circumflex over (V)}_(o), while theswitches in the series tuned version of an AC processing pickup circuitsuch as that shown in FIG. 2 must be rated to handle the peak resonantinductor voltage, along with the peak load current, Î_(o).

If a DC output voltage is required to be derived from the parallel tunedcircuit of FIG. 1 the peaks of the AC voltage input to the rectifier(not shown) must be 157% of the output DC voltage. The same is true forthe current in series tuned circuit of FIG. 2 when a DC output isrequired. As the circuit Q, output power, and output voltage increase,the limitations of the available switch technology become increasinglydifficult to design around with both standard parallel and series tunedAC processing topologies.

In the parallel tuned, AC Processing pickup described in WO2010/030195A1, the AC switch is directly in parallel with the resonantinductor and must therefore be rated to survive the peak resonantinductor current and peak load voltage. Therefore, assuming that thecircuit is designed to run such that Q_(operational)=Q₂, theV_(sw)·I_(sw) or VA product which the switch must be rated for is√{square root over (2)}V_(oc)Q_(z)·√{square root over (2)}I_(sc)√{squareroot over (Q²+1)}, which is slightly above 2V_(oc)I_(sc)Q₂ ². However,the purpose of the AC switch is to control the circuit Q. If more thanI_(sc) is allowed to flow through the switch in parallel with the ACload, the circuit Q falls. If less than I_(sc) is allowed to flowthrough the switch in parallel with the AC load, the circuit Q rises. Assuch, the minimum VA rating of the switch must be at least the shortcircuit current, multiplied by the pickup's maximum output voltage(I_(sc)Q₂V_(oc)). Therefore, the VA rating of both the switches is >2Q₂times the minimum switch rating required for power control.

It would be advantageous to provide a circuit topology for which therequired peak switch voltage and current ratings can be reduced.

OBJECT OF THE INVENTION

It is an object of the present invention to provide apparatus or methodsin connection for pick-up circuits in connection with pick-up circuitsfor IPT systems which reduce switch stresses. Alternatively, it is anobject of the present invention to at least provide a useful alternativeto known apparatus or methods.

SUMMARY OF THE INVENTION

In one aspect the invention provides an inductive power transfer (IPT)pick-up circuit for receiving power from a primary conductor at aselected operating frequency, the circuit having:

-   -   a pick-up coil and a compensation capacitor;    -   a switch means controllable between an operable state and an        inoperable state;    -   a plurality of additional reactive elements whereby when the        switch means is in one of the operable or inoperable states the        additional reactive elements are resonant at the selected        operating frequency to substantially reduce or prevent power        being supplied to an output of the pick-up circuit.

Preferably the switch means is in the other of the operable orinoperable states power is supplied to the load. Preferably the switchmeans is in the other of the operable or inoperable states the pick-upcoil and compensation capacitor become resonant.

Preferably, when the switch means is in the other of the operable orinoperable states the pick-up coil, compensation capacitor and at leastone of the plurality of reactive elements are together resonant.

In one embodiment the pick-up coil and compensation capacitor areconnected in parallel. Preferably the output is provided in parallelwith the compensation capacitor.

Preferably the plurality of additional reactive components comprise acapacitor and inductor connected in series.

Preferably the series connected capacitor and inductor are connected inparallel with the pick-up coil and compensation capacitor.

Preferably the switch means is connected in parallel with the capacitor.Preferably the output includes a rectifier to provide a DC power supply.

In one embodiment the pick-up coil and compensation capacitor arearranged in series. Preferably the output is provided in series with thecompensation capacitor.

Preferably the plurality of additional reactive components comprise acapacitor and inductor connected in parallel. Preferably the parallelconnected capacitor and inductor are connected in series with thepick-up coil and compensation capacitor.

Preferably the switch means is connected in series with the inductor.Preferably an additional compensation capacitor is connected in parallelwith the inductor to reduce a peak switch voltage.

Preferably the output includes a rectifier to provide a DC power supply.

In a further aspect the invention provides an IPT system including apick-up circuit according to any one of the preceding statements.

In a further aspect the invention provides a method in controlling aninductive power transfer (IPT) pick-up circuit having a pick-up coil anda compensation capacitor, the method including the steps of operating aswitch means to cause one or more additional reactive components tobecome resonant to thereby control the power supply to a load.

Preferably the method includes disposing the switch means in one of anoperable or inoperable state to cause the reactive elements to beresonant to substantially prevent power being supplied to an output ofthe pick-up circuit.

Preferably the method includes disposing the switch means in the otherof the operable or inoperable states to cause power to the load.

Preferably the method includes disposing the switch means in the otherof the operable or inoperable states to cause the pick-up coil andcompensation capacitor to become resonant to supply power to the load.

In another aspect the invention provides an inductive power transfer(IPT) pick-up circuit for receiving power from a primary conductor at aselected operating frequency, the circuit having:

-   -   a pick-up coil;    -   a tuning capacitor connected in series with the pick-up coil;    -   a switch means controllable between an operable state and an        inoperable state to control the transfer of power to the pick-up        circuit;    -   a compensation capacitor connected in parallel with the pick-up        coil to reduce a peak switch voltage.

Preferably the compensation capacitor is non-resonant with the pick-upcoil at the operation frequency when the switch is in the inoperablestate. Preferably the switch means is connected in series with thepick-up coil.

Preferably a load supplied by the pick-up circuit is connected in serieswith the pick-up coil.

Further aspects of the invention will become apparent from the followingdescription.

BRIEF DESCRIPTION OF THE DRAWINGS

One or more examples of the invention will be discussed below withreference to the accompanying drawings in which:

FIG. 1: is a circuit diagram of a known parallel tuned IPT pick-upcircuit.

FIG. 2: is a circuit diagram of a known series tuned IPT pick-upcircuit.

FIG. 3: is a circuit diagram of an improved parallel tuned IPT pick-upcircuit.

FIG. 4: is a diagram of the pick-up circuit of FIG. 3 with a DC output.

FIG. 5: Shows operating wave forms for the circuit of FIG. 4.

FIG. 6. Shows operating wave forms for the circuit of FIG. 3.

FIG. 7: Shows measured load current (upper trace) and voltage (lowertrace) of a 1.2 kilowatt system according to FIG. 1 operating at threedifferent controlled output voltage levels:

-   -   (a). 50 volts;    -   (b). 120 volts;    -   (c). 240 volts.

FIG. 8: Shows measured load current (upper trace) and voltage (lowertrace) of a 1.2 kilowatt system according to FIG. 3 operating at threedifferent controlled output voltage levels:

-   -   (a). 50 volts;    -   (b). 120 volts;    -   (c). 240 volts.

FIG. 9: Shows design of normalised switch current rating (relative tothe pick-up short circuit current) by setting the circuit ratioK_(L)=L₂/L₃.

FIG. 10: Shows design of normalised switch voltage rating under variousoperating Q's (relative to the pick-up open circuit voltage) by settingthe circuit ratio K_(L)=L₂/L₃. (a) K_(L)=1, (b) K_(L)=5.

FIG. 11: Shows operating waveforms of the circuit of FIG. 3 with (a)I_(L2) (upper trace) and I_(L3) (lower trace), (b) output voltage (uppertrace) and switch voltage (lower trace)

FIG. 12: Shows a circuit diagram of a new series tuned IPT pick-upcircuit.

FIG. 13: Shows the circuit of FIG. 12 including a DC output.

FIG. 14: Operating waveforms of the circuit of FIG. 12.

FIG. 15: Shows design conditions for the circuit of FIG. 12 withselection of capacitor ratio K_(c)=C₃/C₂ on (a) normalised switchcurrent, (b) normalised switch voltage

FIG. 16: Shows series tuned resonant circuits of FIGS. 2 and 12 can usepartial parallel tuning capacitor via an additional C_(2p) to minimiseswitch transients. Here it is shown applied to the circuit of FIG. 2.

FIG. 17: Shows the circuit topology of FIG. 12 but including anon-resonant parallel capacitor for reducing peak switch voltages.

FIG. 18: Shows the circuit topology of FIG. 13 but including anon-resonant parallel capacitor for reducing peak switch voltages.

FIG. 19: Shows total harmonic distortion comparison between differentparallel tuned AC regulators and a standard Commercial Dimmer for alighting load.

The diagrams shown in FIGS. 5-11, 14 and 15 have been derived from afundamental model view of the circuits to which they relate. As such,the waveforms may be slightly different in practical circuits which arelikely to experience harmonics from switching transients.

DESCRIPTION OF ONE OR MORE EMBODIMENTS OF THE INVENTION

Two new controller topologies are proposed herein which lower switchstress. One is applicable for the parallel tuned cases of the ACProcessing Controller described in WO 2010/030195A1. The other isapplicable to series tuned circuits such as that of FIG. 2 which isdescribed in WO 2011/046453. When applied to meet realistic IPT systemrequirements with a resonant tuning factor of Q₂=5, the new topologiesresult in switch VA ratings falling by up to 3.8 times compared to thetraditional AC processing regulators. Under the proposed topologies,switch blocking voltages are controllable by the designer, independentof the required output voltage. For Q₂ values above 5, the relativeswitch stress will fall further.

The AC output voltage from the proposed parallel and series resonantcontrollers are also sinusoidal, unlike in the AC pickup of WO2010/030195A1 and WO 2011/046453. As such, RFI and harmonic distortionacross the load will also reduce.

The proposed parallel tuned resonant controller operates at zero outputpower when the AC switch is off, whereas the proposed series tunedresonant controller operates at maximum power when the AC switch is off,which in this topology improves its efficiency over that of FIG. 2.

Both topologies can be configured to enable a controlled DC output,although the new series-tuned resonant controller naturally results in asmaller and more cost effective DC output pickup design than is possiblewith the standard parallel tuned pickup for lower voltage DC outputbecause a DC inductor is not required.

A Modified AC Controlled Parallel-Tuned Resonant Controller

To regulate the proposed pickup circuit's output power using practicalswitches with the lowest loss, variations on the previously discussed ACprocessing circuits are herein proposed which do not expose the switchto such high voltages and currents. One example of such a circuit,applicable to parallel tuned pickup coils, is shown in FIG. 3. Thoseskilled in the art will understand that there are potentially manydifferent arrangements which reduce switch stress but only one exampleis discussed here so that the concepts may be clearly understood andwill be referred to in the remainder of this document as the “paralleltuned, resonant controller”. This controller is based upon the ACswitch, composed of S₁, S₂, D₁ and D₂ in FIG. 3, and uses the method ofclamping the resonant capacitor (C₃ in this case) to change theimpedance of the adjacent inductor-capacitor pair (L₃ and C₃) asdescribed in WO 2010/030195A1.

The following description explains how the circuit of FIG. 3 functions.Here L₂ is a pickup inductor which is loosely coupled to an IPT primaryconductor such as a track (not shown) operating at a frequency in theVLF to LF range (typically 10-140 kHz for medium to high powerapplications). The pickup inductor L₂ is tuned to resonate at the trackfrequency using compensation capacitor C₂ and L₃ as described furtherbelow, so that when L₂ is resonant, power is transferred from theprimary conductor to the pick-up circuit.

If L₃ and C₃ in FIG. 3 are additional reactive elements that are tunedto resonance at the track frequency, it can be seen that, when the ACswitch is left as an open circuit, L₃ and C₃ will resonate and form ashort circuit across the pickup inductor L₂. If this effective shortcircuit occurs, the power delivered to the load (R₂) will be zero. Ifthe compensation capacitor C₂ in FIG. 3 is chosen such that it resonateswith the combined parallel impedance of L₂ and L₃ at the track frequencyit can be observed that, if the AC switch is short circuited, thecombined reactance of L₃, C₃ and the AC switch, as seen by the parallelresonant tank, will simply be X_(L3) (i.e. the reactance of L₃). Thus,the power delivered to the load will be at maximum because the combinedparallel impedance of C₂, L₂ and L₃ is resonant at the track frequency.Therefore, when the AC switch is in an operable (i.e. closed) state,power is supplied to the load. When the AC switch is in an inoperable(i.e. open) state, no power is transferred from the primary conductor,so power is not supplied to the load, or at least the supply of power tothe load (i.e. the output) is substantially reduced or prevented.

If a DC output is desired then the circuit can be modified to that shownin FIG. 4 by adding a rectifier across the tuning capacitor C₂ alongwith an output filter in the form of inductor L_(DC) and C_(DC).

The simulated operation of the circuit of FIG. 4 is shown in FIG. 5,where the output power is varied by varying the time that each of theswitches are on in each half cycle of the track frequency (T/2). Thistime for which the switches remain on is called the clamp time (T_(c))which can theoretically vary from 0-T/2. In this simulation the trackfrequency is set to 20 kHz so that 25 pts represents a theoreticalmaximum clamp time. If the clamp time is zero there is no powerdelivered to the load, while with full clamp time the maximum power isavailable to the load. As shown in FIG. 5, T_(c) is initially at amaximum and is slowly reduced to near zero and then ramped back to closeto maximum, enabling power to be varied. Initially full power is output,after which the controller ramps down to half power, then ramps down tozero power, and then ramps back up to full power. While the output powercan be increased rapidly without detriment, if the output power isdecreased rapidly without control, high transient switch voltages willresult. It should be noted that ramping the voltage with suitable slewrate eliminates this problem as shown in FIG. 5, and for the purposes oflighting control such a slew rate enables controlled reduction in lightoutput much faster than would be possible with lighting controllersoperating at 50 Hz mains frequency.

A particular advantage of the proposed parallel-tuned resonantcontroller is that it operates at zero output power when the switchesare off, meaning that during the start up period when the IPT system isfirst turned on, the controller will have as much time as it requires tostart-up, without the resonant voltage building up uncontrollably. It isalso naturally safe if the controller fails, since it will naturallydecouple the pickup when the switches are in the off state. Furthermore,the output voltage in the proposed parallel-tuned, resonant controlleris perfectly sinusoidal, unlike in the standard parallel tuned ACProcessing Controller of FIG. 1. This means peak load voltages are lowerfor a given RMS value and that RFI is reduced. FIG. 6 shows that thevoltage across the load is sinusoidal, even though the voltage acrossthe switches is distorted due to the clamping control. FIG. 7 and FIG. 8show the difference in operation of known controller of FIG. 1 and theproposed controlled of FIG. 3 respectively driving a 1.2 kW Phillips240V stage light which is highly resistive operating from a commonpickup that has a V_(oc)=85V and I_(sc)=6 A. As shown in FIG. 7, whenoperated from the circuit of FIG. 1 the switch voltage and output ACvoltage are necessarily the same and there is notable distortion in thevoltage and current due to the clamping action of the switch. FIG. 8shows operation with identical output RMS voltages across this lightunder operation of the circuit of FIG. 3. As shown there is nonoticeable distortion in either the voltage or current.

A practical drawback of the proposed parallel-tuned, resonant controllercircuit, of FIG. 3, is the trade off between inductor L₃'s volume,denoted here as proportional to G_(L) ₃ , (where G_(L) ₃ =L₃/I_(L) ₃ ²)and the blocking voltage rating required of the switch. Consider thecase where the circuit is delivering full output power. Due to the ACswitch, C₃ is fully shorted, so the equivalent circuit consists of L₂,L₃, C₂ and R₂ in parallel. Therefore, I_(L) ₃ =V_(o)/X_(L) ₃=V_(o)/(ωL₃). As such G_(L) ₃ =V_(o) ²/(ω²L₃) and given V_(o) is loaddependent while ω is typically fixed depending on the requirements ofthe IPT design (making both largely beyond the control of the designer),G_(L) ₃ can be reduced by increasing L₃. However, as L₃ increases,simulation shows that V_(C) _(s) (and therefore the switch voltageV_(sw)) also increases. The ratio of maximum I_(L) _(s) to maximum I_(L)₂ can be chosen by design using the ratio:

L _(L)=(V _(o) /X _(L) ₃ )/V _(o) /X _(L) ₂ )=L ₂ /L ₃.

The impact that K_(L) has on the switch current and voltage is shown inFIGS. 9 and 10 respectively. FIG. 9 shows the switch current ratingnormalised relative to the pick-up short circuit current for variousvalues of K_(L) at fixed Q. FIG. 10 shows the switch voltage ratingnormalised relative to the pick-up open circuit voltage for variousvalues of Q. Here FIG. 10(a) has a K_(L)=1 whereas FIG. 10(b) has avalue K_(L)=5. As shown the maximum steady state value is given(independent of operating Q) as V_(sw)=K_(L)V_(oc), whereas from FIG. 9where Q=3, it is shown that the steady state switch currentI_(sw)=(Q/K_(L))I_(sc). As such K_(L) is useful to ensure the switchvoltage and current can be matched to suitably available switches for agiven application. FIG. 11 shows controlled operation of this newcircuit of FIG. 3 when K_(L)=3˜Q_(2max), and it is providing controlledoutput to a 1.2 kW light. As expected the switch currents remain similarto I_(sc).

By way of example, if a circuit based on the DC output topology of FIG.4 was required and designed to meet the following parameters: f=20 kHz,P_(su)=V_(oc)I_(sc)=400 VA, P_(o)=2 kW at 400V DC output, then assuminga known practical pick-up has an L₂=157 μH, V_(oc)=88.9V and I_(sc)=4.5A, it can be calculated from these specifications that the requiredQ₂=5. If L₃ is set to equal to L₂, the LI² rating of L₃ will equal thatof L₂. According to simulation, the peak switch voltage will then be˜166V, the peak switch current will be 32.5 A and the total VA rating ofthe switch is at least 5.4 kVA. Practical high power switches normallyhave better voltage blocking capability than current ratings, so adesign that requires a low voltage but high current switch isundesirable. The switch blocking voltage (V_(DS)) and the resonantvoltage across the pickup coil (V_(C2)) are shown in FIG. 4, at avariety of output power levels. The peak switch current is the same asthe peak current flowing through I_(L3). By changing the design so thatL₃=L₂ Q₂, the peak switch voltage changes to approximately the peak ACload voltage, which is what it would be in the traditional AC Processingpickup circuit of FIG. 1. However while the peak switch voltage hasrisen to 648V, the peak switch current falls to 8.6 A, and the volume ofL₃ falls to 2.5 times less than that of L₂. The VA rating of the switchis now 5.6 kVA, similar to the last example.

To meet the same parameters using the conventional parallel tuned, ACprocessing circuit of FIG. 1 requires a switch voltage rating of atleast √{square root over (2)}{circumflex over (V)}_(c) ₂ =√{square rootover (2)}(π/2√{square root over (2)})V_(DC) or 628V. The peak switchcurrent rating would need to be √{square root over (2)}I_(sc)√{squareroot over (Q²+1)}, or 32.4 A. Therefore, the total VA rating of theswitch must be at least 20.3 kVA, or approximately 3.8 times higher thanwhen using the resonant controller of FIG. 4. Consequently, the VArating of the switch for the newly proposed circuit has fallensignificantly and the desired switch voltage can now be setindependently of the output voltage by suitable choice of the inductorratio: K_(L)=L₂/L₃.

A Modified AC Controlled Series Tuned Resonant Controller

The series tuned AC Processing controller shown in FIG. 2 as describedin WO 2011/046453 can also be modified to reduce the peak switch voltageand current requirements. An example of such a modified controller isshown in FIG. 12, here referred to as a “series tuned, resonantcontroller”.

The following description explains how the circuit of FIG. 12 functions.Here L₂ is a pickup inductor which is loosely coupled to an IPT primaryconductor such as a track (not shown) operating at a frequency in theVLF to LF range (typically 10-140 kHz for medium to high powerapplications). The pickup inductor L₂ is tuned to resonate at the trackfrequency using compensation capacitor C₂ and C₃ as described furtherbelow, so that when L₂ is resonant power is transferred from the primaryconductor to the pick-up circuit.

Inductor L₃ and capacitor C₃ are additional reactive components. Supposethat inductor L₃ is chosen to resonate with C₃ at the track frequency.Consider the case where the AC switch, composed of S₁, S₂, D₁ and D₂, ison. Because C₃ and L₃ form a parallel resonant circuit at the trackfrequency, they will together appear as an open circuit to the rest ofthe circuit, thereby reducing the power delivered to the load to zero.Now suppose that capacitor C₂ is chosen such that it resonates at thetrack frequency when in series with capacitor C₃ and pickup inductor L₂.When the AC switch is off, the inductor L₃ will be disconnected from therest of the circuit, and because C₂, C₃ and the pickup inductor L₂ areresonant at the track frequency, the power delivered to the load is at amaximum. Therefore, in this embodiment, when the AC switch is in aninoperable (i.e. open) state, power is supplied to the load. When the ACswitch is in an operable (i.e. closed) state, no power is transferredfrom the primary conductor, so power is not supplied to the load, or atleast the supply of power to the load (i.e. the output) is substantiallyreduced or prevented.

As in the parallel tuned resonant controller proposed earlier, a DCoutput can be produced from this circuit, in this case by adding arectifier followed by a DC capacitor in place of the AC load. Becausethis is a series tuned system the DC inductor is not required. This DCoutput arrangement is shown in FIG. 13.

A circuit based on the proposed series-tuned, resonant controllerpowering a 220V, 1200 W AC load can be designed and simulated. Becauseof the transiently high current required by tungsten-halogenincandescent lamps as they turn on, series tuned IPT pickups may be moreapplicable to driving these loads than parallel tuned circuits. However,because the peak switch voltage in the standard series tuned ACProcessing pickup is √{square root over (2)}V_(oc)√{square root over (Q₂²+1)}, the standard series tuned AC Processing topology of FIG. 2 is notparticularly suited to applications where a high output voltage isrequired. The modified circuit of FIG. 12 however overcomes thisproblem.

By way of example, if a series tuned circuit is designed to power astage light to meet the following specifications: f=20 kHz Pickup coilP_(Su)=240 VA, P_(o)=1.2 kW at 220V AC output using a pickup inductorL2=1.61 mH that has a Voc=220V, Isc=1.09 A, then using thesespecifications, it can be determined that Q₂=5. According to simulation,if C₃=QC₂, the LI² rating of inductor L₃ will be 3.6 times smaller thanthat of the pickup inductor coil, L₂. In consequence L₃ will have avolume˜3.6 times less than L₂. According to simulation, the results ofwhich are shown in FIG. 14, the peak switch voltage will be 386V, andthe peak switch current will be 9.5 A. Therefore, the total VA rating ofthe switch must be at least 3.7 kVA. In practice the ratio K_(c)=C₂/C₃can be chosen to adjust the maximum switch current required under steadystate operation as shown in FIG. 15(a), without affecting the switchvoltage (FIG. 15(b)). IN RMS terms I_(sw)=(K_(c)+1)I_(sc) however thepeak switch current is Î_(sw)=√{square root over (2)}(K_(c)+1)I_(sc).

To meet the same parameters using the conventional series-tuned ACProcessing pickup circuit of FIG. 2 the peak switch voltage would be atleast √{square root over (2)}V_(oc)√{square root over (Q₂ ²+1)}=1.59 kV.The switch current rating would need to be >Q₂I_(sc)=5.45 A. Therefore,the total VA rating of the switch must be at least 8.7 kVA, or 2.4 timeshigher than if the series-tuned resonant controller of FIG. 12 is used.Switches capable of blocking 1.59 kV while operating at the VLFfrequencies as required for IPT applications are not usually practical,as such the standard series tuned AC processing circuit of FIG. 2 is notsuited to meet the proposed specifications. However, as in the paralleltuned case, the series tuned resonant controller of FIG. 12 may besuitable. FIG. 12 operates at full power when the switch is off, whereaswhen the switch is on, the switch current is relatively small.Consequently the proposed pickup's efficiency should be higher than thestandard series tuned AC Processing pickup of FIG. 2.

As with the proposed parallel-tuned resonant controller, there is atrade off between peak switch voltage and inductor L₃'s volume and cost.However, when using the series-tuned resonant controller topology for DCoutput as shown in the circuit of FIG. 13, several hundred volts outputmay be produced without requiring excessive switch blocking voltages.Because a DC inductor is not required to produce DC from the seriestuned pickup, any additional size and cost in L₃ may be partially orcompletely offset by the lack of a DC inductor.

Minimising Transients in the Series Resonant Circuits

As stated above, the output voltage of FIG. 12 is perfectly sinusoidal,unlike that of the standard parallel tuned AC processing controller ofFIG. 2. This means peak load voltages are lower for a given RMS valueand RFI is reduced. Switch voltage oscillation and RFI is a particularproblem for the standard series tuned AC processing circuit and needs tobe addressed here as well. The turn off of the standard series tuned ACcontroller (FIG. 2) occurs when the body diode of the off-state switchreverse biases. At this point, the reverse recovery current of the diodestarts to flow through the pickup inductor coil L₂ but rapidly falls tozero as the reverse recovery charge is depleted. This large δi/δtthrough the pickup inductor causes a positive voltage spike across theblocking half of the AC switch. The voltage spike then induces highfrequency oscillation in the switch blocking voltage V_(DS) as theswitch's capacitance C_(DS) resonates with the pickup inductor. As wellas creating EMI, this increases the peak voltage present across theswitches significantly. A simple RC snubber across both of the switchescan remedy this problem however the snubber's resistive losses reducethe pickup's efficiency.

A novel lossless approach is to parallel tune the pickup inductor with amarginal amount of voltage boost before the series tuning capacitor, asshown in FIG. 16. This gives the reverse recovery current a lowimpedance path across the pickup inductor coil, minimizing voltagespikes and oscillations as the AC switch turns off. Simulation has shownthat this acts as an effective snubber by minimizing the peak switchvoltages. The impedance of the proposed additional parallel compensationcapacitor (C_(2P)) (which is non-resonant with the pick-up coil L₂ atthe operation frequency when the AC switch is off i.e. open orinoperable) as used in simulation was set to −10X_(L2), resulting in avoltage boost of 10%. In practice the addition of this capacitor changesthe effective inductance of L₂ and this must be taken into account inthe design of the other tuning components in the circuit. For aC_(2P)=−10X_(L2), the effective new secondary inductance (L₂′) whichtakes into account the presence of C_(2P) is L₂′=1.1L₂.

The switch blocking voltage oscillation problem outlined here alsoapplies to L₃ in the proposed series-tuned resonant controller of FIG.12, however, the problem can also be mitigated by the use of theadditional capacitor, allowing additional freedom in switch selectionand consequently lower switch voltage and VA requirements.

FIGS. 17 and 18 show the topologies of FIGS. 12 and 13 respectively, butwith an additional compensation capacitor C_(L3P) in parallel withinductor L₃. As described in the preceding paragraph, C_(L3P) provides apath for the current through the inductor L₃ when the AC switch isopened and thus avoids the need for a snubber. For example, thenon-resonant parallel capacitor C_(L3P) may have a reactance that isapproximately ten times that of the inductor L₃, so that the combinedreactance of these parallel components is inductive and approximately110% of the reactance of L₃.

From the foregoing it can be seen that the invention providessignificant advantages in circuit control and efficiency and inminimising switch stresses. Furthermore, although the discussion abovefocuses on controlling the pick-up circuit using an AC switch incombination with one of more reactive elements to achieve a controlledvariable reactive element, other configurations are possible. Forexample configurations which generate a variable reactance are alsoknown within the field of inductive transfer, such as using a “saturableinductor/reactor” or switching a binary weighted capacitor bank usingrelays for example. An inductor/capacitor tuning branch as described inthis document could be applied to these other configurations as well,and may result in similar or the same advantages as when tuning thecircuit with an AC switch.

The new circuits also considerably harmonic distortion across the load.An example is shown in FIG. 19 which shows the level of total harmonicdistortion for given output power for a lighting load. Line 1 representsa standard commercial dimmer circuit. Line 2 represents use of aparallel tuned pick-up circuit using the topology of FIG. 1. Line 3,which shows a very low level of distortion, represents use of new theparallel tuned pick-up topology disclosed herein, such as that of FIG.3.

Moreover, the person skilled in the art will appreciate that whereas theknown circuit described in WO 2010/030195 reflects the capacitive loadon the track, the circuits described in this document may reflect aninductive load onto the track. These reactive loads may be used tocontrol tuning aspects of the overall IPT system. In one example,different pick-up circuit topologies may be used in separate pick-ups toachieve a desired result whereby the reflected reactances effectivelytune each other out.

Those skilled in the art will also appreciate that, as with the circuitdescribed in WO 2010/030195, the new circuits disclosed herein alsoallow the Q of the circuit to be controlled and/or fine tuned.

It will be seen from the foregoing that various changes to modificationsto the presently preferred embodiments described in herein will beapparent to those skilled in the art. Such changes in modifications maybe made without departing the spirit and scope of the present inventionand without dimensioning its intended advantages. It is therefore,intended, that such changes and modifications be included within thepresent invention.

1. An inductive power transfer (IPT) pick-up circuit for receiving powerfrom a primary conductor at a selected operating frequency, the circuitcomprising: a pick-up coil and a compensation capacitor; a switchcontrollable between an operable state and an inoperable state; aplurality of additional reactive elements whereby when the switch is inone of the operable or inoperable states the additional reactiveelements are resonant at the selected operating frequency tosubstantially reduce or prevent power being supplied to an output of thepick-up circuit.
 2. An IPT pick-up circuit as claimed in claim 1 whereinwhen the switch is in the other of the operable or inoperable statespower is supplied to the load.
 3. An IPT pick-up as claimed in claim 1wherein when the switch is in the other of the operable or inoperablestates the pick-up coil and compensation capacitor become resonant. 4.An IPT pick-up as claimed in claim 1 wherein when the switch is in theother of the operable or inoperable states the pick-up coil,compensation capacitor and at least one of the plurality of reactiveelements are together resonant.
 5. (canceled)
 6. An IPT pick-up circuitas claimed in claim 1 wherein the pick-up coil and compensationcapacitor are connected in parallel.
 7. An IPT pick-up circuit asclaimed in claim 6 wherein the output is provided in parallel with thecompensation capacitor.
 8. An IPT pick-up circuit as claimed in claim 6wherein the plurality of additional reactive components comprise acapacitor and inductor connected in series.
 9. An IPT pick-up circuit asclaimed in claim 8 wherein the series connected capacitor and inductorare connected in parallel with the pick-up coil and compensationcapacitor.
 10. An IPT pick-up circuit as claimed in claim 9 wherein theswitch means is connected in parallel with the capacitor.
 11. (canceled)12. An IPT pick-up circuit as claimed in claim 1 wherein the pick-upcoil and compensation capacitor are arranged in series.
 13. An IPTpick-up circuit as claimed in claim 12 wherein the output is provided inseries with the compensation capacitor.
 14. An IPT pick-up circuit asclaimed in claim 12 wherein the plurality of additional reactivecomponents comprise a capacitor and inductor connected in parallel. 15.An IPT pick-up circuit as claimed in claim 14 wherein the parallelconnected capacitor and inductor are connected in series with thepick-up coil and compensation capacitor. 16-20. (canceled)
 21. A methodin controlling an inductive power transfer (IPT) pick-up circuit havinga pick-up coil and a compensation capacitor, the method comprisingoperating a switch to cause one or more additional reactive componentsto become resonant to thereby control the power supply to a load.
 22. Amethod as claimed in claim 21 further comprising disposing the switch inone of an operable or inoperable state to cause the reactive elements tobe resonant to substantially prevent power being supplied to an outputof the pick-up circuit.
 23. A method as claimed in claim 22 furthercomprising disposing the switch in the other of the operable orinoperable states to control power to the load.
 24. A method as claimedin claim 23 further comprising disposing the switch in the other of theoperable or inoperable states to cause the pick-up coil and compensationcapacitor to become resonant to supply power to the load.
 25. Aninductive power transfer (IPT) pick-up circuit for receiving power froma primary conductor at a selected operating frequency, the circuitcomprising: a pick-up coil; a tuning capacitor connected in series withthe pick-up coil; a switch controllable between an operable state and aninoperable state to control transfer of power to the pick-up circuit; acompensation capacitor connected in parallel with the pick-up coil toreduce a peak switch voltage.
 26. An IPT pick-up circuit as claimed inclaim 25 wherein the compensation capacitor is non-resonant with thepick-up coil at the selected operating frequency when the switch is inthe inoperable state. 27-30. (canceled)